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| april 2005 1 m9999-040805 mic2168 micrel, inc. mic2168 1mhz pwm synchronous buck control ic general description the mic2168 is a high-ef?ciency, simple to use 1mhz pwm synchronous buck control ic housed in a small msop-10 package. the mic2168 allows compact dc/dc solutions with a minimal external component count and cost. the mic2168 operates from a 3v to 14.5v input, without the need of any additional bias voltage. the output voltage can be precisely regulated down to 0.8v. the adaptive all n-channel mosfet drive scheme allows ef?ciencies over 95% across a wide load range. the mic2168 senses current across the high-side n-chan - nel mosfet, eliminating the need for an expensive and lossy current-sense resistor. current limit accuracy is main - tained by a positive temperature coef?cient that tracks the increasing r ds(on) of the external mosfet. further cost and space are saved by the internal in-rush-current limiting digital soft-start. the mic2168 is available in a 10-pin msop package, with a wide junction operating range of C40c to +125c. all support documentation can be found on micrels web site at www.micrel.com. typical application 1.2h 3.3v v in = 5v vdd comp/en vin cs fb gnd lsd bst 1k? 10k? 4k? 3.24k? 4.7f 100f 0.1f 100nf irf7821 sd103bws irf7821 100pf hsd vsw mic2168 150f x 2 mic2168 adjustable output 1mhz converter features ? 3v to 14.5v input voltage range ? adjustable output voltage down to 0.8v ? up to 95% ef?ciency ? 1mhz pwm operation ? adjustable current-limit senses high-side n-channel mosfet current ? no external current sense resistor ? adaptive gate drive increases ef?ciency ? ultra-fast response with hysteretic transient recovery mode ? overvoltage protection protects the load in fault conditions ? dual mode current limit speeds up recovery time ? hiccup mode short-circuit protection ? internal soft-start ? dual function comp and en pin allows low-power shut - down ? small size msop 10-lead package applications ? point-of-load dc/dc conversion ? set-top boxes ? graphic cards ? lcd power supplies ? telecom power supplies ? networking power supplies ? cable modems and routers micrel, inc. ? 2180 fortune drive ? san jose, ca 95131 ? usa ? tel + 1 (408) 944-0800 ? fax + 1 (408) 474-1000 ? http://www.micrel.com
mic2168 micrel, inc. m9999- 040805 2 april 2005 pin con?guration fb gnd 6 5 1 vin vdd cs comp/en 10 bst hsd vsw lsd 9 8 7 2 3 4 10-pin msop (mm) pin description pin number pin name pin function 1 vin supply voltage (input): 3v to 14.5v. 2 vdd 5v internal linear regulator (output): v dd is the external mosfet gate drive supply voltage and an internal supply bus for the ic. when v in is <5v, this regulator operates in dropout mode. 3 cs current sense / enable (input): current-limit comparator noninverting input. the current limit is sensed across the mosfet during the on time. the cur - rent can be set by the resistor in series with the cs pin. 4 comp/en compensation (input): dual function pin. pin for external compensation. if this pin is pulled below 0.2v, with the reference fully up the device shuts down (50 a typical current draw). 5 fb feedback (input): input to error ampli?er. regulates error ampli?er to 0.8v. 6 gnd ground (return). 7 lsd low-side drive (output): high-current driver output for external synchro - nous mosfet. 8 vsw switch (return): high-side mosfet driver return. 9 hsd high-side drive (output): high-current output-driver for the high-side mosfet. when v in is between 3.0v to 5v, 2.5v threshold-rated mosfets should be used. at v in > 5v, 5v threshold mosfets should be used. 10 bst boost (input): provides the drive voltage for the high-side mosfet driver. the gate-drive voltage is higher than the source voltage by v in minus a diode drop. ordering information part number frequency junction temp. range package standard pb-free MIC2168BMM mic2168ymm 1mhz - 40 c to +125 c 10 -lead msop april 2005 3 m9999-040805 mic2168 micrel, inc. absolute maximum ratings (1) supply voltage (v in ) ................................................... 15.5v booststrapped voltage (v bst ) ................................. v in +5v junction temperature (t j ) ................... C40c t j +125c storage temperature (t s ) ........................ C65c to +150c operating ratings (2) supply voltage (v in ) ..................................... +3v to +14.5v output voltage range .......................... 0.8v to v in d max package thermal resistance ja 10-lead msop ............................................. 180c/w electrical characteristics (3) t j = 25c, v in = 5v, unless otherwise speci?ed. bold values indicate C40c < t j < +125c parameter condition min typ max units feedback voltage reference ( 1%) 0.792 0.8 0.808 v feedback voltage reference ( 2% over temp) 0.784 0.8 0.816 v feedback bias current 30 100 na output voltage line regulation 0.03 % / v output voltage load regulation 0.5 % output voltage total regulation 3v v in 14.5v; 1a i out 10a; (v out = 2.5v) (4) 0.6 % oscillator section oscillator frequency 900 1000 1100 khz maximum duty cycle 90 % minimum on-time (4) 30 60 ns input and v dd supply pwm mode supply current v cs = v in C0.25v; v fb = 0.7v (output switching but excluding 1.6 3 ma external mosfet gate current.) shutdown quiescent current v comp/en = 0v 50 150 a v comp shutdown threshold 0.1 0.25 0.4 v v comp shutdown blanking c comp = 100nf 4 ms period digital supply voltage (v dd ) v in 6v 4.7 5 5.3 v notes: 1. absolute maximum ratings indicate limits beyond which damage to the component may occur. electrical speci?cations do not apply when operating the device outside of its operating ratings. the maximum allowable power dissipation is a function of the maximum junction temperature, t j (max), the junction-to-ambient thermal resistance, ja , and the ambient temperature, t a . the maximum allowable power dissipation will result in excessive die temperature, and the regulator will go into thermal shutdown. 2. devices are esd sensitive, handling precautions required. 3. speci?cation for packaged product only. 4. guaranteed by design. mic2168 micrel, inc. m9999- 040805 4 april 2005 parameter condition min typ max units error ampli?er dc gain 70 db transconductance 1 ms soft-start soft-start current after timeout of internal timer. see soft-start section. 8.5 a current sense cs over current trip point v cs = v in C0.25v 160 200 240 a temperature coef?cient +1800 ppm/c output fault correction thresholds upper threshold, v fb_ovt (relative to v fb ) +3 % lower threshold, v fb_uvt (relative to v fb ) C3 % gate drivers rise/fall time into 3000pf at v in > 5v 30 ns output driver impedance source, v in = 5v 6 ? sink, v in = 5v 6 ? source, v in = 3v 10 ? sink, v in = 3v 10 ? driver non-overlap time note 6 10 20 ns notes: 5. speci?cation for packaged product only. 6. guaranteed by design. electrical characteristics (5) april 2005 5 m9999-040805 mic2168 micrel, inc. typical characteristics v in = 5v mic2168 micrel, inc. m9999- 040805 6 april 2005 functional description the mic2168 is a voltage mode, synchronous step-down switching regulator controller designed for high output power without the use of an external sense resistor. it includes an internal soft-start function which reduces the power supply input surge current at start-up by controlling the output volt - age rise time, a pwm generator, a reference voltage, two mosfet drivers, and short-circuit current limiting circuitry to form a complete 1mhz switching regulator. theory of operation the mic2168 is a voltage mode step-down regulator. the ?gure above illustrates the block diagram for the voltage control loop. the output voltage variation due to load or line changes will be sensed by the inverting input of the transcon - ductance error ampli?er via the feedback resistors r3, and r2 and compared to a reference voltage at the non-invert - ing input. this will cause a small change in the dc voltage level at the output of the error ampli?er which is the input to the pwm comparator. the other input to the comparator is a 0 to 1v triangular waveform. the comparator generates a rectangular waveform whose width t on is equal to the time from the start of the clock cycle t 0 until t 1 , the time the triangle crosses the output waveform of the error ampli?er. to illustrate the control loop, let us assume the output volt - age drops due to sudden load turn-on, this would cause the inverting input of the error ampli?er which is divided down functional diagram current limit reference current limit comparator error amp low-side driver high-side driver pwm comparator fb comp gnd lsd v ref +3% v ref 3% hsd v dd c bst cs v dd 5v 5v 5v c2 c1 r1 5v 0.8v v in sw q2 q1 l1 driver logic 0.8v bg v ali d clamp & startup current enable error loop hys comparator 5v ldo bandgap reference soft-start & digital delay counter mic2168 ramp clock boost r2 r3 4w rsw rcs c out v out c in d1 mic2168 block diagram version of v out to be slightly less than the reference voltage causing the output voltage of the error ampli?er to go high. this will cause the pwm comparator to increase t on time of the top side mosfet, causing the output voltage to go up and bringing v out back in regulation. soft-start the comp/en pin on the mic2168 is used for the following three functions: 1. disables the part by grounding this pin 2. external compensation to stabilize the voltage control loop 3. soft-start for better understanding of the soft-start feature, lets as - sume v in = 12v, and the mic2168 is allowed to power-up by un-grounding the comp/en pin. the comp pin has an internal 8.5 a current source that charges the external com - pensation capacitor. as soon as this voltage rises to 180mv (t = cap_comp 0.18v/8.5 a), the mic2168 allows the internal v dd linear regulator to power up and as soon as it crosses the undervoltage lockout of 2.6v, the chips internal oscillator starts switching. at this point in time, the comp pin current source increases to 40 a and an internal 11-bit counter starts counting which takes approximately 2ms to complete. during counting, the comp voltage is clamped at 0.65v. after this counting cycle the comp current source april 2005 7 m9999-040805 mic2168 micrel, inc. is reduced to 8.5 a and the comp pin voltage rises from 0.65v to 0.95v, the bottom edge of the saw-tooth oscillator. this is the beginning of 0% duty cycle and it increases slowly causing the output voltage to rise slowly. the mic2168 has two hysteretic comparators that are enabled when v out is within 3% of steady state. when the output voltage reaches 97% of programmed output voltage, then the g m error ampli?er is enabled along with the hysteretic comparator. this point onwards, the voltage control loop (g m error ampli?er) is fully in control and will regulate the output voltage. soft-start time can be calculated approximately by adding the following four time frames: t1 = cap_comp 0.18v/8.5 a t2 = 12 bit counter, approx 2ms t3 = cap_comp 0.3v/8.5 a v v out in t 4 ? ? ? ? ? ? x 0.5 - x cap_comp 8.5 a soft-start time(cap_comp=100nf) = t1 + t2 + t3 + t4 = 2.1ms + 2ms + 3.5ms + 1.8ms = 10ms current limit the mic2168 uses the r ds(on) of the top power mosfet to measure output current. since it uses the drain to source resistance of the power mosfet , it is not very accurate. this scheme is adequate to protect the power supply and external components during a fault condition by cutting back the time the top mosfet is on if the feedback voltage is greater than 0.67v. in case of a hard short when feedback voltage is less than 0.67v, the mic2168 discharges the comp capacitor to 0.65v, resets the digital counter and automatically shuts off the top gate drive, and the g m error ampli?er and the C3% hysteretic comparators are completely disabled and the soft-start cycles restarts. this mode of operation is called the hiccup mode and its purpose is to protect the down stream load in case of a hard short. the circuit in figure 1 illustrates the mic2168 current limiting circuit. l1 inductor v in hsd lsd rcs cs 200a 0 c2 c in c1 c out q1 mosfe t n q2 mosfe t n v out figure 1. the mic2168 current limiting circuit the current limiting resistor r cs is calculated by the follow - ing equation: r cs = 200 a r ds(on) q1 l l equation (1) i l = 2(inductor ripple current ) 1 i load = where: inductor ripple current = (v in v out = - v out ) v in f switching l f switching = 1mhz 200 a is the internal sink current to program the mic2168 current limit. the mosfet r ds(on) varies 30% to 40% with temperature; therefore, it is recommended to add a 50% margin to the load current (i load ) in the above equation to avoid false current limiting due to increased mosfet junction temperature rise. it is also recommended to connect r cs resistor directly to the drain of the top mosfet q1, and the r sw resistor to the source of q1 to accurately sense the mosfets r ds(on) . a 0.1 f capacitor in parallel with r cs should be connected to ?lter some of the switching noise. internal v dd supply the mic2168 controller internally generates v dd for self bias - ing and to provide power to the gate drives. this v dd supply is generated through a low-dropout regulator and generates 5v from v in supply greater than 5v. for supply voltage less than 5v, the v dd linear regulator is approximately 200mv in dropout. therefore, it is recommended to short the v dd supply to the input supply through a 10 ? resistor for input supplies between 2.9v to 5v. mosfet gate drive the mic2168 high-side drive circuit is designed to switch an n-channel mosfet. the block diagram in figure 2 shows a bootstrap circuit, consisting of d2 and cbst, supplies energy to the high-side drive circuit. capacitor cbst is charged while the low-side mosfet is on and the voltage on the vsw pin is approximately 0v. when the high-side mosfet driver is turned on, energy from cbst is used to turn the mosfet on. as the mosfet turns on, the voltage on the vsw pin increases to approximately v in . diode d2 is reversed biased and cbst ?oats high while continuing to keep the high-side mosfet on. when the low-side switch is turned back on, cbst is recharged through d2. the drive voltage is derived from the internal 5v v dd bias supply. the nominal low-side gate drive voltage is 5v and the nominal high-side gate drive voltage is approximately 4.5v due the voltage drop across d2. an approximate 20ns delay between the high- and low-side driver transitions is used to prevent current from simultane - ously ?owing unimpeded through both mosfets. mosfet selection the mic2168 controller works from input voltages of 3v to 13.2v and has an internal 5v regulator to provide power to turn the external n-channel power mosfets for high- and low-side switches. for applications where v in < 5v, the internal mic2168 micrel, inc. m9999- 040805 8 april 2005 v dd regulator operates in dropout mode, and it is necessary that the power mosfets used are low threshold and are in full conduction mode for v gs of 2.5v. for applications when v in > 5v; logic-level mosfets, whose operation is speci?ed at v gs = 4.5v must be used. it is important to note the on-resistance of a mosfet increases with increasing temperature. a 75c rise in junction tempera - ture will increase the channel resistance of the mosfet by 50% to 75% of the resistance speci?ed at 25c. this change in resistance must be accounted for when calculating mosfet power dissipation and in calculating the value of curren t -sense (cs) resistor. total gate charge is the charge required to turn the mosfet on and off under speci?ed operating conditions (v ds and v gs ). the gate charge is supplied by the mic2168 gate drive circuit. at 1mhz switching frequency and above, the gate charge can be a signi?cant source of power dissipation in the mic2168. at low output load, this power dissipation is noticeable as a reduction in ef?ciency. the average current required to drive the high-side mosfet is: i q f g[high - side](avg) g s = where: i g[high-side](avg) = average high-side mosfet gate current. q g = total gate charge for the high-side mosfet taken from manufacturers data sheet for v gs = 5v. the low-side mosfet is turned on and off at v ds = 0 because the freewheeling diode is conducting during this time. the switching loss for the low-side mosfet is usu - ally negligible. also, the gate-drive current for the low-side mosfet is more accurately calculated using ciss at v ds = 0 instead of gate charge. for the low-side mosfet: i c v f g[low - side](avg) iss gs s = since the current from the gate drive comes from the input voltage, the power dissipated in the mic2168 due to gate drive is: p v i i gatedrive in g[high - side](avg) g[low - side](avg) = + ( ) a convenient ?gure of merit for switching mosfets is the on resistance times the total gate charge r ds(on) q g . lower numbers translate into higher ef?ciency. low gate-charge logic-level mosfets are a good choice for use with the mic2168. parameters that are important to mosfet switch selection are: ? voltage rating ? on-resistance ? total gate charge the voltage ratings for the top and bottom mosfet are essentially equal to the input voltage. a safety factor of 20% should be added to the v ds (max) of the mosfets to account for voltage spikes due to circuit parasitics. the power dissipated in the switching transistor is the sum of the conduction losses during the on-time (p conduction ) and the switching losses that occur during the period of time when the mosfets turn on and off (p ac ). p p p sw conduction ac = + where: p i r conduction sw(rms) sw 2 = p p p ac ac(off) ac(on) = + r sw = on-resistance of the mosfet switch d = v v o in ? ? ? ? ? ? duty cycle making the assumption the turn-on and turn-off transition times are equal; the transition times can be approximated by: t c v c v i t iss gs oss in g = + where: c iss and c oss are measured at v ds = 0 i g = gate-drive current (1a for the mic2168) the total high-side mosfet switching loss is: p ( v v ) i t f ac in d p k t s = + where: t t = switching transition time (typically 20ns to 50ns) v d = freewheeling diode drop, typically 0.5v f s it the switching frequency, nominally 1mhz the low-side mosfet switching losses are negligible and can be ignored for these calculations. inductor selection values for inductance, peak, and rms currents are required to select the output inductor. the input and output voltages and the inductance value determine the peak-to-peak induc - tor ripple current. generally, higher inductance values are used with higher input voltages. larger peak-to-peak ripple currents will increase the power dissipation in the inductor and mosfets. larger output ripple currents will also require more output capacitance to smooth out the larger ripple cur - rent. smaller peak-to-peak ripple currents require a larger inductance value and therefore a larger and more expensive inductor. a good compromise between size, loss and cost is to set the inductor ripple current to be equal to 20% of the maximum output current. the inductance value is calculated by the equation below. l v ( v m ax v ) v m ax f 0.2 i max out in out in s out = ? ( ) ( ) ( ) where: f s = switching frequency, 1mhz 0.2 = ratio of ac ripple current to dc output current v in (max) = maximum input voltage the peak-to-peak inductor current (ac ripple current) is: april 2005 9 m9999-040805 mic2168 micrel, inc. i v ( v m ax v ) v m ax f l pp out in out in s = ? ( ) ( ) the peak inductor current is equal to the average output current plus one half of the peak-to-peak inductor ripple current. i i max 0.5 i pk out pp = + ( ) the rms inductor current is used to calculate the i 2 r losses in the inductor. i i ma x 1 1 3 i i m ax inductor(rms) out p out 2 = + ? ? ? ? ? ? ( ) ( ) maximizing ef?ciency requires the proper selection of core material and minimizing the winding resistance. the high frequency operation of the mic2168 requires the use of fer - rite materials for all but the most cost sensitive applications. lower cost iron powder cores may be used but the increase in core loss will reduce the ef?ciency of the power supply. this is especially noticeable at low output power. the winding resistance decreases ef?ciency at the higher output current levels. the winding resistance must be minimized although this usually comes at the expense of a larger inductor. the power dissipated in the inductor is equal to the sum of the core and copper losses. at higher output loads, the core losses are usually insigni?cant and can be ignored. at lower output currents, the core losses can be a signi?cant con - tributor. core loss information is usually available from the magnetics vendor. copper loss in the inductor is calculated by the equation below: p i r inductor cu inductor(rms) winding 2 = the resistance of the copper wire, r winding , increases with temperature. the value of the winding resistance used should be at the operating temperature. r r 1 0 .0042 (t t ) winding(hot) winding(20 c) hot 20 c = + ? ( ) where: t hot = temperature of the wire under operating load t 20c = ambient temperature r winding(20c) is room temperature winding resistance (usu - ally speci?ed by the manufacturer) output capacitor selection the output capacitor values are usually determined capaci - tors esr (equivalent series resistance). voltage and rms current capability are two other important factors selecting the output capacitor. recommended capacitors tantalum, low-esr aluminum electrolytics, and poscaps. the output capacitors esr is usually the main cause of output ripple. the output capacitor esr also affects the overall voltage feedback loop from stability point of view. see feedback loop compensation section for more information. the maximum value of esr is calculated: r v i esr out pp ? where: v out = peak-to-peak output voltage ripple i pp = peak-to-peak inductor ripple current the total output ripple is a combination of the esr output capacitance. the total ripple is calculated below: ? v i ( 1 d ) c f i r out pp out s 2 pp esr 2 = ? ? ? ? ? ? ? + ( ) where: d = duty cycle c out = output capacitance value f s = switching frequency the voltage rating of capacitor should be twice the voltage for a tantalum and 20% greater for an aluminum electrolytic. the output capacitor rms current is calculated below: i i 12 c pp out(rms) = the power dissipated in the output capacitor is: p i r diss(c c esr(c ) out out(rms) 2 out ) = input capacitor selection the input capacitor should be selected for ripple current rating and voltage rating. tantalum input capacitors may fail when subjected to high inrush currents, caused by turning the input supply on. tantalum input capacitor voltage rating should be at least 2 times the maximum input voltage to maximize reliability. aluminum electrolytic, os-con, and multilayer polymer ?lm capacitors can handle the higher inrush currents without voltage derating. the input voltage ripple will primarily depend on the input capacitors esr. the peak input current is equal to the peak inductor current, so: ? v i r in inductor(peak) esr(c ) in = the input capacitor must be rated for the input current ripple. the rms value of input capacitor current is determined at the maximum output current. assuming the peak-to-peak inductor ripple current is low: mic2168 micrel, inc. m9999- 040805 10 april 2005 i i ma x d (1 d) c (rms) out in ? ( ) the power dissipated in the input capacitor is: p i r diss(c ) c (rms) esr(c ) in in 2 in = voltage setting components the mic2168 requires two resistors to set the output voltage as shown in figure 2. error amp 7 mic2168 [adj.] fb v ref 0.8v r2 r1 figure 2. voltage-divider con?guration where: v ref for the mic2168 is typically 0.8v the output voltage is determined by the equation: v v 1 r1 r2 o ref = + ? ? ? ? ? ? a typical value of r1 can be between 3k ? and 10k ? . if r1 is too large, it may allow noise to be introduced into the voltage feedback loop. if r1 is too small, in value, it will decrease the ef?ciency of the power supply, especially at light loads. once r1 is selected, r2 can be calculated using: r2 v r 1 v v ref o ref = ? external schottky diode an external freewheeling diode is used to keep the inductor current ?ow continuous while both mosfets are turned off. this dead time prevents current from ?owing unimpeded through both mosfets and is typically 15ns. the diode conducts twice during each switching cycle. although the average current through this diode is small, the diode must be able to handle the peak current. the reverse voltage requirement of the diode is: v v diode(rrm) in = the power dissipated by the schottky diode is: p i v diode d(avg) f = where: v f = forward voltage at the peak diode current the external schottky diode, d1, is not necessary for circuit operation since the low-side mosfet contains a parasitic body diode. the external diode will improve ef?ciency and decrease high frequency noise. if the mosfet body diode is used, it must be rated to handle the peak and average cur - rent. the body diode has a relatively slow reverse recovery time and a relatively high forward voltage drop. the power lost in the diode is proportional to the forward voltage drop of the diode. as the high-side mosfet starts to turn on, the body diode becomes a short circuit for the reverse recovery period, dissipating additional power. the diode recovery and the circuit inductance will cause ringing during the high-side mosfet turn-on. an external schottky diode conducts at a lower forward voltage preventing the body diode in the mosfet from turning on. the lower forward voltage drop dissipates less power than the body diode. the lack of a reverse recovery mechanism in a schottky diode causes less ringing and less power loss. depending on the circuit components and operating conditions, an external schottky diode will give a 1 / 2 % to 1% improvement in ef?ciency. feedback loop compensation the mic2168 controller comes with an internal transcon - ductance error ampli?er used for compensating the voltage feedback loop by placing a capacitor (c1) in series with a resistor (r1) and another capacitor c2 in parallel from the comp pin to ground. see functional block diagram. power stage the power stage of a voltage mode controller has an induc - tor, l1, with its winding resistance (dcr) connected to the output capacitor, c out , with its electrical series resistance (esr) as shown in figure 3. the transfer function g(s), for such a system is: esr c out v o dcr l figure 3. the output lc filter in a voltage mode buck converter (1 + esr s c) ? ? ? ? ? ? - g (s) dcr s c + s 2 l c + 1 + esr s c plotting this transfer function with the following assumed values (l=2 h, dcr=0.009 ? , c out =1000 f, esr=0.050 ? ) gives lot of insight as to why one needs to compensate the loop by adding resistor and capacitors on the comp pin. figures 4 and 5 show the gain curve and phase curve for the above transfer function. april 2005 11 m9999-040805 mic2168 micrel, inc. 100 1 . 1 0 3 1 . 1 0 4 1 . 1 0 5 1 . 1 0 6 60 37.5 15 7.5 30 30 60 gain 10000 00 100 f figure 4. the gain curve for g(s) 100 1 . 10 3 1 . 10 4 1 . 10 5 1 . 10 6 150 100 50 0 0 180 p hase 1000 000 100 f figure 5. phase curve for g(s) it can be seen from the transfer function g(s) and the gain curve that the output inductor and capacitor create a two pole system with a break frequency at: 1 = f c 2 l c out therefore, f lc = 3.6khz by looking at the phase curve, it can be seen that the output capacitor esr (0.050 ? ) cancels one of the two poles (lc out ) system by introducing a zero at: 1 = f zero 2 esr c out therefore, f zero = 6.36khz. from the point of view of compensating the voltage loop, it is recommended to use higher esr output capacitors since they provide a 90 phase gain in the power path. for comparison purposes, figure 6, shows the same phase curve with an esr value of 0.002 ? . 100 1 . 10 3 1 . 10 4 1 . 10 5 1 . 10 6 150 100 50 0 0 180 p hase 1000 000 100 f figure 6. the phase curve with esr = 0.002 ? it can be seen from figure 5 that at 50khz, the phase is approximately C90 versus figure 6 where the number is C150. this means that the transconductance error ampli - ?er has to provide a phase boost of about 45 to achieve a closed loop phase margin of 45 at a crossover frequency of 50khz for figure 4, versus 105 for figure 6. the simple rc and c2 compensation scheme allows a maximum error ampli?er phase boost of about 90. therefore, it is easier to stabilize the mic2168 voltage control loop by using high esr value output capacitors. g m error ampli?er it is undesirable to have high error ampli?er gain at high frequencies because high frequency noise spikes would be picked up and transmitted at large amplitude to the output, thus, gain should be permitted to fall off at high frequencies. at low frequency, it is desired to have high open-loop gain to attenuate the power line ripple. thus, the error ampli?er gain should be allowed to increase rapidly at low frequencies. the transfer function with r1, c1, and c2 for the internal g m error ampli?er can be approximated by the following equation: 1+ r1 s c1 g m s (c1 + c2) error amplifier (z) - ? ? ? ? ? ? 1 + r1 c1 c2 s c1 + c2 the above equation can be simplified by assuming c2< april 2005 13 m9999-040805 mic2168 micrel, inc. design example layout and checklist: 1. connect the current limiting (cs) resistor directly to the drain of top mosfet q1. 2. connect the vsw pin directly to the source of top mosfet q1 thru a 4 ? to 10 ? resistor. the purpose of this resistor is to ?lter the switch node. 3. the feedback resistors r1 and r2 should be placed close to the fb pin. the top side of r1 should connect directly to the output node. run this trace away from the switch node (junction of q1, q2, and l1). the bottom side of r1 should connect to the gnd pin on the mic2168. 4. the compensation resistor and capacitors should be placed right next to the comp/en pin and the other side should connect directly to the gnd pin on the mic2168 rather than going to the plane. 5. the input bulk capacitors should be placed close to the drain of the top mosfet. 6. the 1 f ceramic capacitor should be placed right on the vin pin of the mic2168. 7. the 4.7 f to 10 f ceramic capacitor should be placed right on the vdd pin. 8. the source of the bottom mosfet should connect directly to the input capacitor gnd with a thick trace. the output capacitor and the input capacitor should connect directly to the gnd plane. 9. place a 0.1 f ceramic capacitor in parallel with the cs resistor to ?lter any switching noise. mic2168 micrel, inc. m9999- 040805 14 april 2005 package information r e v . 0 0 10-pin msop (mm) micrel inc. 2180 fortune drive san jose, ca 95131 usa tel + 1 (408) 944-0800 fax + 1 (408) 474-1000 web http://www.micrel.com this information furnished by micrel in this data sheet is believed to be accurate and reliable. however no responsibility is assumed by micrel for its use. micrel reserves the right to change circuitry and speci?cations at any time without noti?cation to the customer. micrel products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product can reasonably be expected to result in personal injury. life support devices or systems are devices or systems that (a) are intended for surgical implant into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a signi? cant injury to the user. a purchaser's use or sale of micrel products for use in life support appliances, devices or systems is a purchaser's own risk and purchaser agrees to fully indemnify micrel for any damages resulting from such use or sale. ? 2003 micrel incorporated |
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